Scalable periphery tunable matching power amplifier

ABSTRACT

A scalable periphery tunable matching power amplifier is presented. Varying power levels can be accommodated by selectively activating or deactivating unit cells of which the scalable periphery tunable matching power amplifier is comprised. Tunable matching allows individual unit cells to see a constant output impedance, reducing need for transforming a low impedance up to a system impedance and attendant power loss. The scalable periphery tunable matching power amplifier can also be tuned for different operating conditions such as different frequencies of operation or different modes.

CROSS REFERENCE TO RELATED APPLICATIONS—CLAIM OF PRIORITY

The present application is a continuation of commonly owned co-pendingU.S. patent application Ser. No. 15/827,984 filed on Nov. 30, 2017 andis hereby incorporated by reference herein in its entirety. ApplicationSer. No. 15/827,984 is a continuation of commonly owned U.S. patentapplication Ser. No. 14/957,399 filed on Dec. 2, 2015 (now U.S. Pat. No.9,847,759 issued Dec. 19, 2017), and is also hereby incorporated byreference herein in its entirety. Application Ser. No. 14/957,399 is acontinuation of commonly owned U.S. patent application Ser. No.13/797,779 filed on Mar. 12, 2013 (now U.S. Pat. No. 9,294,056 issuedMar. 22, 2016), and is also hereby incorporated by reference herein inits entirety.

Patent application Ser. No. 13/797,779 may be related to U.S. patentapplication Ser. No. 13/797,686 filed on Mar. 12, 2013 (now U.S. Pat.No. 9,602,063 issued Mar. 21, 2017), entitled “Variable Impedance Matchand Variable Harmonic Terminations for Different Modes and FrequencyBands”, which is incorporated herein by reference in its entirety.

BACKGROUND 1. Field

The present disclosure relates to power amplifiers. More particularly,the present disclosure relates to scalable periphery tunable matchingpower amplifiers.

2. Description of Related Art

In the field of mobile radios, a typical power amplifier vendor isdependent on the ability to quickly change either a power amplifier (PA)or a power amplifier module (PAM). Power amplifiers and power amplifiermodules are typically needed to meet custom specifications from mobileradio manufacturers to meet their customer's system specifications. Thiscan be a difficult task for those PAs and PAMs implemented in silicon(Si) technologies due in part to long design cycle times and also longfabrication cycle times. These Si technologies include, but are notlimited to, complementary metal oxide semiconductor (CMOS) technology,silicon on insulator (SOI) CMOS technology, silicon on sapphire (SOS)CMOS technology, and bipolar CMOS (BiCMOS) technology (e.g. technologyinvolving a combination of bipolar junction transistors and CMOStransistors).

A power amplifier is an important component in many radio frequency (RF)systems, as the power amplifier is usually necessary to amplify an RFsignal prior to transmitting the RF signal using an antenna. Duringcellular voice communications, a power amplifier typically spends asmall fraction of operation time operating at peak power level. However,during wireless data communications, the power amplifier can spend asignificant portion of operation time operating at peak power level toallow more efficient data transmission.

Various cellular communications devices operate on different frequencybands in the RF spectrum. For example, there are frequency bands in useat 700-900 MHz and 1700-2400 MHz as well as many others. Currentcellular devices are not capable of covering multiple bandssimultaneously because of expense incurred when adding extra componentsthat would enable such cellular devices to cover multiple bandssimultaneously. However, an amplifier that can be tuned can be utilizedto enable a power amplifier in an RF front-end that covers many bands.For this reason, tunability of power amplifiers shows promise for futurepower amplifier design.

SUMMARY

According to a first aspect of the present disclosure, an amplifier isprovided, the amplifier comprising: a scalable periphery amplifiercomprising one or more unit cells connected in parallel with each otherand adapted to be selectively activated or deactivated, wherein eachunit cell comprises a plurality of active devices configured to operateas an amplifier; an output tunable matching network operativelyconnected to an output of the scalable periphery amplifier, wherein thetunable matching network is configured to adjust an output loadimpedance seen by the output of the scalable periphery amplifier; and anamplifier control circuitry configured to selectively activate ordeactivate the one or more unit cells, thus varying a total output powerfrom the amplifier, wherein the plurality of active devices of the oneor more unit cells are a stack of a plurality of transistors, an inputtransistor of the plurality of transistors in the stack being configuredto receive an input signal of the scalable periphery amplifier.

According to a second aspect of the present disclosure, an amplifier isprovided, the amplifier comprising: a scalable periphery amplifiercomprising one or more unit cells connected in parallel with each otherand adapted to be selectively activated or deactivated, wherein at leastone unit cell comprises one transistor configured to operate as anamplifier; an output tunable matching network operatively connected toan output of the scalable periphery amplifier, wherein the tunablematching network is configured to adjust an output load impedance seenby the output of the scalable periphery amplifier; and an amplifiercontrol circuitry configured to selectively activate or deactivate theone or more unit cells, thus varying a total output power from theamplifier, wherein the one transistor of the at least one unit cell isconfigured to receive an input signal of the scalable peripheryamplifier.

According to a third aspect of the present disclosure, a method forbiasing a scalable periphery amplifier is provided, the methodcomprising: providing a plurality of amplifiers of the scalableperiphery amplifier, wherein each amplifier of the plurality ofamplifiers of the scalable periphery amplifier comprises a plurality oftransistors configured in a stacked arrangement; operatively couplingthe plurality of amplifiers in a parallel arrangement; providing aninput power range for an input signal to the plurality of amplifiers ofthe scalable periphery amplifier; providing a desired power gain of thescalable periphery amplifier; obtaining an expected output power rangeof the scalable periphery amplifier based on the desired power gain andthe input power range; based on the obtaining, activating one or moreamplifiers from the plurality of amplifiers.

According to a fourth aspect of the present disclosure, a method foramplifying an input signal is provided, the method comprising: providinga plurality of amplifiers, wherein each amplifier of the plurality ofamplifiers comprises a plurality of transistors configured in a stackedarrangement; operatively coupling the plurality of amplifiers in aparallel arrangement; providing a plurality of switches configured toselectively activate or deactivate the plurality of amplifiers; couplinga tunable matching network to an input port of the plurality ofamplifiers; inputting an input signal to the tunable matching network;measuring the input signal power level; activating one or moreamplifiers from the plurality of amplifiers based on the input signalpower level and on a desired output signal power level, and adjusting animpedance of the tunable matching network in correspondence of an inputsignal power level shift in correspondence of further activating ordeactivating of the one or more amplifiers from the plurality ofamplifiers, wherein the further activating or deactivating of the one ormore amplifiers is in correspondence of a desired output signalcharacteristic.

BRIEF DESCRIPTION OF DRAWINGS

FIG. 1 shows a vector representation of a modulated RF signal in thebaseband [I, Q] plane.

FIG. 2 shows a signal constellation for the case of 16-QAM.

FIG. 3 shows an exemplary class A amplifier comprised of an FET with itsaccompanying current I_(D) and voltage V_(D) characteristics.

FIG. 4 shows current and voltage characteristics for a class A amplifieroperating with reduced signal input.

FIG. 5 shows voltage and current characteristics of an amplifier when aDC/DC converter is used to shift a DC bias point of the amplifier inresponse to reduced signal input.

FIG. 6 shows voltage and current characteristics of an amplifier whenslope of a load line is adjusted to have a steeper downward slope(corresponding to a lower load resistance) together with reducing drainvoltage in response to a reduced signal input.

FIG. 7A shows current and voltage characteristics of an amplifier whenbias current is reduced.

FIG. 7B shows current and voltage characteristics of an individualactive unit cell of the amplifier of FIG. 7A. Note that the scale ofcurrent on the y-axis is shown for the unit cell, not the full sizestructure of FIG. 7A.

FIG. 8A shows current and voltage characteristics of an amplifier whenbias current is reduced and load impedance seen by the entire poweramplifier is increased.

FIG. 8B shows current and voltage characteristics of an individualactive unit cell of the amplifier of FIG. 8A. Note that the scale ofcurrent on the y-axis is shown for the unit cell, not the full sizestructure of FIG. 8A.

FIGS. 9A-9C show an exemplary scalable periphery tunable matching poweramplifier architecture comprising a scalable periphery amplifier and atunable matching network.

FIG. 10A shows an exemplary arrangement of unit cells of the scalableperiphery amplifier shown in FIGS. 9A-9C.

FIGS. 10B-10D show embodiments of a unit cell of the scalable peripheryamplifier of FIGS. 9A-9C.

FIG. 10E shows an exemplary embodiment wherein a capacitor at a gate ofa transistor and a bias voltage adjustment allow for equalization ofvoltages across the stack of transistors.

FIG. 10F shows an exemplary embodiment wherein the stack of transistorscomprises N-type and P-type transistors.

FIG. 10G shows an alternative embodiment of FIG. 10A wherein a couplingcapacitor in the RF input path to the amplifier is not integrated withineach unit cell.

FIG. 11 shows an exemplary tunable matching network that can be utilizedin the SPTM power amplifier architecture of FIG. 9A-9C.

FIG. 12 shows an exemplary embodiment of the disclosure comprisingmultiple SPTM amplifiers.

FIGS. 13-14 show a graphical representation of Pout vs. ACLR (adjacentchannel leakage ratio) for class A and class AB operation respectively,when half of the total number of unit cells are on, compared with whenall of the unit cells are on.

FIG. 15 shows PAE (power added efficiency) and ACLR as a function ofoutput power for both class A and class AB operation.

FIG. 16 shows PAE as a function of output power for both class A andclass AB operation.

FIG. 17 shows ACLR as a function of output power for both class A andclass AB operation.

FIG. 18 shows an example implementation of switching between variousclasses of amplifiers.

FIG. 19 shows an exemplary realization of the tunable matching networkof FIG. 11, wherein switches and tunable reactive elements are used.

FIG. 20 shows an example implementation of using variable impedancecircuits to affect a harmonic termination for a desired class ofoperation.

DETAILED DESCRIPTION

As used in the present disclosure, the terms “switch ON” and “activate”may be used interchangeably and can refer to making a particular circuitelement electronically operational.

As used in the present disclosure, the terms “switch OFF” and“deactivate” may be used interchangeably and can refer to making aparticular circuit element electronically non-operational.

As used in the present disclosure, the terms “amplifier” and “poweramplifier” may be used interchangeably and can refer to a device that isconfigured to amplify a signal input to the device to produce an outputsignal of greater magnitude than the magnitude of the input signal.

As used in the present disclosure, the term “mode” can refer to awireless standard and its attendant modulation and coding scheme orschemes. As different modes may require different modulation schemes,these may affect required channel bandwidth as well as affect thepeak-to-average-ratio (PAR), also referred to aspeak-to-average-power-ration (PAPR), as well as other parameters knownto the skilled person. Examples of wireless standards include GlobalSystem for Mobile Communications (GSM), code division multiple access(CDMA), Worldwide Interoperability for Microwave Access (WiMAX), LongTerm Evolution (LTE), as well as other wireless standards identifiableto a person skilled in the art. Examples of modulation and codingschemes include binary phase-shift keying (BPSK), quadrature phase-shiftkeying (QPSK), quadrature amplitude modulation (QAM), 8-QAM, 64-QAM, aswell as other modulation and coding schemes identifiable to a personskilled in the art.

As used in the present disclosure, the term “efficiency” can refer to ameasure of how well a device converts one energy source to another. In acase of an NMOS device configured as a power amplifier, (drain)efficiency of the NMOS device is a metric that quantifies conversion ofdirect current (DC) power that is input to a drain of the NMOS device toRF power output from the drain of the NMOS device. Power addedefficiency (PAE) provides a metric of efficiency that takes intoconsideration that gain of the power amplifier is not infinite.Corresponding drain efficiency and PAE equations are provided in a latersection.

As used in the present disclosure, the term “channel” can refer to afrequency range. More in particular, the term “channel” as used hereinrefers to a frequency range that can be defined by a wireless standardsuch as, but not limited to, wideband code division multiple access(WCDMA) and long term evolution (LTE). As is well known, adjacentchannel leakage ratio (ACLR) provides a ratio of adjacent channel powerto main channel power. For instance, in WCDMA, a channel bandwidth is 5MHz. If power is to be transmitted on/within a main channel of 1925 MHzto 1930 MHz, adjacent channels would encompass 5 MHz below the mainchannel (i.e. 1920 MHz to 1925 MHz) and 5 MHz above the main channel(i.e. 1930 MHz to 1935 MHz), and the ACLR will be ratio of power leakedinto one or the other adjacent channel to power in the main channel(e.g. ACLR (low) will refer to a ratio using the power leaked into thelower adjacent channel). Consequently, the ACLR can be utilized as ameasure of linearity of a device.

According to several embodiments of the present disclosure, scalableperiphery, tunable matching, digital control circuitry, and/or otheradjustments can be utilized to reconfigure a power amplifier afterfabrication (e.g. at final test, where product personality plusmanufacturing offsets can be stored in the power amplifier) or duringoperation of a device containing the power amplifier. A need toreconfigure a power amplifier may be in response to a change in acustomer's specification or to enable use of a single PA platform fordifferent applications that span different frequencies, different powerranges, and/or different modes.

If techniques of scalable periphery (SP) and tunable matching (TM) areapplied together in accordance with several embodiments of the presentdisclosure, a PA platform can be tuned, for example, for multiplewireless standards and/or multiple frequency bands. This means that asingle PA with scalable periphery and tunable matching can be used inmultiple products without a new design cycle, which also translates toreduced manufacturing and production cost. Furthermore, a given wirelessdevice containing a single PA with scalable periphery and tunablematching can be compatible with many different frequency bands and/ordifferent wireless standards. Technical details are provided in thefollowing discussion.

FIG. 1 shows a vector representation of a modulated RF signal in an [I,Q] plane. A modulated RF signal may be represented in terms of anin-phase component I and a quadrature component Q. A person skilled inthe art should be aware that the quadrature component of a signal is 90degrees (one quarter of a cycle) out of phase compared to the in-phasecomponent. As a result, the in-phase component I usually represents acomponent, either constant or time-varying, multiplied by cos(ω_(c)t)whereas the quadrature component Q usually represents a component,either constant or time-varying, multiplied by sin(ω_(c)t), where ω_(c)represents radian frequency of a carrier wave.

Amplitude of the RF signal is a function of length of a vector drawnfrom the origin to a point representing the RF signal in the [I, Q]plane, and square of the amplitude of the RF signal is proportional topower of the signal. Consequently, if a multiplicity of possiblepositions of the modulated RF signal's representation in the [I, Q]plane forms a circle centered on the origin, then the length of thevector drawn from the origin to any of these possible positions remainsconstant and similarly the power of the RF signal remains constant. Asignal of this type would be referred to as a constant envelope signal,an example of which is a frequency modulated signal. For an exemplarycase of a constant envelope signal, a power amplifier that is configuredto amplify such a signal has no linearity requirements and can remainsaturated.

However, to increase data rate given a fixed bandwidth, it may becomenecessary to transmit signals whose representations in the [I, Q] planeare at differing distances from the origin, implying differentamplitudes and therefore different power levels, to be transmitted. FIG.2 shows a signal constellation for the case of 16-QAM. A person skilledin the art will be aware that other variations of QAM can be used aswell (e.g. 64-QAM or 1024-QAM, among others), or even other digitalmodulation schemes (e.g. PAM, ASK, AM-PSK, etc. . . . ). Because ofdiffering power levels being transmitted and changing at the rate of themodulation, there is now a linearity requirement imposed on the poweramplifier.

Generally, the power amplifier is designed to operate within a specificfrequency range of operation (also referred to as a frequency channel orjust channel). Nonlinearities in a power amplifier can result inspectral distortion, leading to signal leakage into adjacent channelscorresponding to frequencies neighboring those of a main channel withinwhich operation is desired. A power amplifier is generally designed soas to maintain a low ACLR, where a low ACLR signifies lower nonlinearityof the power amplifier. The ACLR may be measured as, for instance, aratio of the power measured in the adjacent RF channel to thetransmitted power in the main channel. More in particular, the powerscan be measured after a receiver filter that is present in an RFreceiver circuit, where the receiver filter comprises a bandpass filterconfigured to reject image signals and otherwise reduce interferencebetween channels. However, the linearity requirement conflicts withpower considerations because higher efficiency typically occurs when thepower amplifier is driven into nonlinear operation. Thus, there is atrade-off between linearity and efficiency.

Another consideration in power amplifier design is a ratio of peaktransmitted power to average transmitted power to a load, a measureoften referred to as peak to average power ratio (PAPR). Normally, it isdesirable to keep the PAPR as low as possible to maximize efficiency. Ingeneral, a PA designer does not have much choice in this matter as PAPRis defined by a system standard (e.g. WCDMA, LTE, and so on) and databeing transmitted. The PA may reduce PAPR as the PA is driven to operatenear compression. Compression is often defined at the 1 dB compressionpoint. Examples of loads to which power may be transmitted include, butare not limited to, an antenna of a cell phone; downstream splitters,cables, or feed network(s) used in delivering cable television serviceto a consumer; a next amplifier in an RF chain at a cellular basestation; or a beam forming network in a phased array radar system. Otherexample loads identifiable by a person skilled in the art can also beutilized.

Additionally, a power amplifier may be required to operate under manydifferent conditions. By way of example, the different conditions may bea result of a change in operating frequency (e.g. frequency of thecarrier wave), a change in modulation scheme (e.g. from 64-QAM to QPSK),or a change in average power level transmitted, a change in batteryvoltage, and/or a change in temperature. For instance, fourth generation(4G) devices transmit signals corresponding to 64, 16, or 4 points inthe [I, Q] plane, corresponding to 64-QAM, 16-QAM, or QPSK, respectivelyfor each subcarrier, and transmit many carriers in an OFDMA (OrthogonalFrequency Division Multiple Access) system. Although one particular modeof operation of the power amplifier can be optimized, attempting todesign the power amplifier for many different modes of operation maylead to degradation in performance metrics such as linearity (e.g. asmeasured using ACLR) and/or efficiency. A mode of operation depends onmodulation and coding scheme, frequency, bandwidth, PAR, and othercharacteristics such as those mentioned previously.

FIG. 3 shows an exemplary class A amplifier comprised of a field effecttransistor (FET) with its accompanying current I_(D) and voltage V_(D)characteristics. When varying an input signal voltage applied to a gateof the FET, the output (drain) operating point of the FET moves along aload line, which is so named because a negative reciprocal of its slopeis equal to a resistive value of a load impedance seen by the drain ofthe FET. A larger amount of power is delivered to the load impedancewhen the output voltage swings from peaks near a breakdown voltage tovalleys at a knee voltage V_(KNEE). Below the knee voltage, theamplifier is operating in the triode region and is not operatingoptimally as an amplifier. A person skilled in the art will notice thatan amount of power equal to a product of the knee voltage and the DCbias current is normally lost.

As previously mentioned, a power amplifier may be evaluated based onefficiency of the power amplifier. By way of example, efficiency can beevaluated based on drain efficiency and/or power added efficiency (PAE).For discussion purposes, the term “efficiency” is used to refer to“drain efficiency”. Drain efficiency can be calculated as a ratio ofsignal output power P_(RF) of the amplifier to direct current (DC) powerP_(DC) fed to the drain of the amplifier. Signal output power P_(RF) iscalculated as a product of root mean square (RMS) voltage V_(RFRMS) andRMS current I_(RFRMS). DC power P_(DC) is calculated as a product ofvoltage V_(BIAS) and current I_(BIAS) at a DC bias point.

With reference to FIG. 3, drain efficiency can be interpretedgraphically as a ratio of one eighth of the area of the cross-hatchedbox to the area of the solid box shown in FIG. 3, as described in detailas follows. Width of the cross-hatched box is equal to twice themagnitude of the output (drain) voltage waveform, and height of thecross-hatched box is equal to twice the magnitude of the output (drain)current waveform. Power is calculated using RMS values of the voltageand current waveforms, which are equal to magnitude values divided bythe square root of two. The combination of these factors leads to thefactor of 8 in the denominator of the expression for drain efficiencyshown in FIG. 3. As described above, equations for drain efficiency,signal power, DC power, RMS voltage, and RMS current can be given by,respectively:Drain Efficiency=P _(RF) /P _(DC)P _(RF) =V _(RFRMS) ×I _(RFRMS)P _(DC) =I _(BIAS) ×V _(BIAS)V _(RFRMS) =V _(RFpk-pk)/(2*sqrt(2))I _(RFRMS) =I _(RF) pk-pk/(2*sqrt(2))Alternatively, if P_(RFin) denotes the signal input power, then the PAEequation is given by:PAE=(P _(RF) −P _(RFin))/P _(DC)

By definition, a class A amplifier conducts current during 100% of asignal cycle of the input signal voltage V_(RF). Although such anamplifier is highly linear, it exhibits lower drain efficiency thanalternative configurations (e.g. class B, class AB, and class C, amongothers). These alternative configurations, which conduct current duringless than 100% of the signal cycle, exhibit nonlinear operation and thusexhibit increased ACLR due to the higher nonlinearity, but generallydemonstrate greater drain efficiency than the class A amplifier.Although example embodiments will be described primarily using anexemplary class A amplifier, the present disclosure can also be usedwith other amplifier classes as well.

FIG. 4 shows current and voltage characteristics for a class A amplifieroperating at the same DC bias point (V_(BIAS), I_(BIAS)) as FIG. 3 butwith reduced signal input relative to FIG. 3. As a result, although theDC power P_(DC) (area of the solid box) remains constant, the signaloutput power P_(RF) (area of the cross-hatched box) has been reduced,leading to reduced drain efficiency.

For a given signal output power (i.e. for a constant area of thecross-hatched box), a technique for enhancing drain efficiency is toreduce DC voltage supplied to the drain of the power amplifier by usinga DC/DC converter. In other words, the voltage value V_(BIAS) of the DCbias point (V_(BIAS), I_(BIAS)) can be reduced. This will shift the DCbias point to the left, reducing the DC power dissipated in the loadconnected to the output (drain) of the amplifier while maintaining thesame signal output power. FIG. 5 shows voltage and currentcharacteristics of an amplifier when a DC/DC converter is used to shiftthe DC bias point in response to reduced signal input.

FIG. 6 shows voltage and current characteristics of an amplifier whenthe slope of the load line is adjusted to have a steeper downward slope(corresponding to a lower load resistance) together with reducing drainvoltage in response to reduced signal input. With reference to the threegraphs shown in FIG. 6, proceeding from the top graph to the middlegraph, the load line is adjusted to have a steeper slope and the DC biasvoltage V_(BIAS) is reduced at the drain of the amplifier. Proceedingfrom the middle graph to the bottom graph, the load line is furtheradjusted to have an even steeper slope and the DC bias voltage V_(BIAS)is further at the drain of the amplifier. Specifically, as shown in thethree graphs of FIG. 6, the slope of the load line can be adjusted tohave a steeper downward slope together with the aforementioned method ofreducing drain voltage, a combination which can increase efficiency at agiven delivered signal output power.

In proceeding from the top graph to the bottom graph of FIG. 6, loadline adjustment is provided by a decrease in output impedance as shownin the steepening downward slope of the load line. In a case where DCbias voltage is reduced, efficiency can be increased through such loadline adjustment because the area of each of the cross-hatched boxes canbe optimized for that given DC bias voltage. For example, going from thetop graph to the middle graph, if the output (drain) DC bias voltageV_(BIAS) were reduced without adjusting the slope of the load line, thecross-hatched rectangle would have been smaller (similar to the graphshown in FIG. 5), corresponding to lower RF power output and thus lowerefficiency relative to the case shown in the middle graph of FIG. 6.

With further reference to the decreasing of output impedance proceedingfrom the top graph to the bottom graph of FIG. 6, the decreased outputimpedance is normally transformed upwards to an impedance (e.g. 50Ω, acommon impedance used in RF circuits) of a system containing the poweramplifier, which can result in increased loss. As is well known, anyimpedance transformation with real components (e.g. resistance due to afinite quality factor) introduces loss. A higher transformation ratiodue to the progressive decreasing of output impedance shown in FIG. 6can result in higher loss relative to a case where the output impedanceis kept constant. Consequently, such an example in FIG. 6 illustrates atradeoff between loss and efficiency.

Instead of reducing DC bias voltage V_(BIAS) to compensate for a reducedinput signal drive, the reduced input signal drive can be compensatedfor by adjusting DC bias current I_(BIAS). According to severalembodiments of the present disclosure, the DC bias current I_(BIAS) canbe adjusted by reducing an effective size of the amplifier (to beexplained as follows). FIG. 7A shows current and voltage characteristicsof an amplifier when bias current I_(BIAS) is reduced. The bias currentI_(BIAS) can be reduced by decreasing power amplifier size, whereadjusting of the power amplifier size can be accomplished by designingthe power amplifier to be comprised of N unit cells operating inparallel (i.e. unit cells see equal voltage) and selectively activating(e.g. turning ON) or deactivating (e.g. turning OFF) a subset of suchunit cells to increase or decrease power amplifier effective size,respectively. Each unit cell can comprise one or more active devices(e.g. NMOSs) that are configured to operate as an amplifier. Forpurposes of discussion, the terms “size” and “effective size” of anamplifier are used interchangeably. The DC power dissipation, due to theproduct of the knee voltage and the DC bias current can be reducedbecause DC bias current is reduced.

If load impedance seen by the entire amplifier remains fixed (i.e.constant slope of the load line), then, for a reduced bias currentI_(BIAS), voltage swing seen by the entire amplifier will be reduced.The reduction in voltage swing is indicated graphically by a smallerwidth of the cross-hatched box as height decreases in FIG. 7A,proceeding from the top graph to the bottom graph. Unit cells of thepower amplifier that remain active (e.g. ON) retain current density;that is, an individual unit cell that remains active will continue toconduct an equal amount of DC bias current. As used in the presentdisclosure, the current density of a unit cell may refer to a biascurrent conducted by a unit cell that is active (e.g. ON), assuming thateach unit cell that is active conducts an equal amount of DC biascurrent.

It is advantageous to keep the bias current in each unit cell fixed asdescribed, rather than decreasing the bias current in a fixed sizeamplifier because a decrease in bias current typically results in adecrease in amplifier's bandwidth capability represented by thetransition frequency (ft) or maximum frequency of oscillation (fmax).The linearity of each unit cell is maintained in this way, as it willsee the same bias conditions. It also keeps the transconductance (gm)constant and preserves the class of amplifier operation.

By way of example and not of limitation, consider an amplifier comprisedof 60 unit cells, where each unit cell conducts 10 mA of DC biascurrent. In a case where all 60 unit cells are active, a total DC biascurrent of 600 mA can be conducted by the amplifier. The case where all60 unit cells are active can correspond to the upper left graph of FIG.7A. If 30 unit cells are deactivated, leaving 30 unit cells active,total DC bias current conducted by the power amplifier decreases to 300mA, but each active unit cell can continue to conduct 10 mA of DC biascurrent. The case where 30 unit cells out of the total 60 unit cellsremain active can correspond to the middle graph of FIG. 7A. If 15 moreunit cells are deactivated, leaving 15 unit cells active, the total DCbias current conducted by the power amplifier decreases to 150 mA. Thecase where 15 unit cells out of the total 60 unit cells remain activecan correspond to the bottom graph of FIG. 7A.

For a constant load impedance and DC bias voltage, the reduced DC biascurrent also results in reduced DC power dissipation as indicated byreduced sizes of the solid box proceeding from the top graph to thebottom graph of FIG. 7A. Because the DC bias current per unit cellremains constant while voltage swing is reduced, individual unit cellsthat remain active see a reduced output impedance when other unit cellsare turned off, as indicated graphically by a steeper slope of the loadline in the graph of FIG. 7B.

It is noted that in some embodiments, unit cells that remain active donot necessarily retain current density; that is, DC bias current in eachunit cell that remains active can vary. A person skilled in the artcould readily perform experiments with varying DC bias current of eachunit cell to achieve desired operating characteristics (e.g. linearity,ACLR, output power, drain current, out of band emissions, Rx band noise,error vector magnitude (EVM), and so forth). For purposes of discussiononly, it is generally assumed that each active unit cell conducts anequal amount of bias current.

FIG. 8A shows current and voltage characteristics of an amplifier whenbias current is reduced and load impedance seen by the entire poweramplifier (i.e. N unit cells taken as a unit rather than individually)is increased. Similar to the discussion with respect to FIG. 7A, size ofthe power amplifier can be adjusted by designing the power amplifier tobe comprised of one or more unit cells operating in parallel (i.e. unitcells see equal voltage) and selectively activating or deactivating asubset of such unit cells to increase or decrease power amplifiereffective size, respectively. Reduction of bias current of the poweramplifier can be achieved through a decreasing of power amplifier size.

The topmost graph of FIG. 8A can correspond to an exemplary case whereall unit cells are active. The middle graph of FIG. 8A can correspond toan exemplary case where some unit cells have been deactivated (thusreducing DC bias current) in combination with increasing load impedanceseen by the entire power amplifier (i.e. N unit cells taken as a unitrather than individually). The bottom graph of FIG. 8A can correspond toan exemplary case where even more unit cells have been deactivated (thusfurther reducing bias current) in combination with further increasingthe load impedance seen by the entire power amplifier. FIG. 8B showscurrent and voltage characteristics present at the drain of one of theunit cells that remains active.

If the load impedance seen by the entire power amplifier is increased,although total DC bias current flowing through the entire poweramplifier is reduced, voltage and current swing of any individual activeunit cell can remain constant. As an example, a typical PA outputimpedance is 3Ω, which is then matched to a load (e.g. 50Ω, a commonimpedance used in RF circuits). Increasing the load impedance at anoutput of the power amplifier, relative to the typical PA outputimpedance of 3Ω, can reduce the transformation ratio, thereby reducingloss. This load impedance adjustment corresponds to a shallower downwardsloping load line than the slope of the load line prior to such loadimpedance adjustment. Through such load impedance adjustment, whiletotal current output of the power amplifier can be reduced bydeactivating (e.g. turning OFF) some unit cells, individual unit cellsthat remain active (e.g. ON) do not see a change in either individualcurrent or voltage swing, as shown in FIG. 8B. The individual unit cellsthat remain active see a constant output impedance as represented by thenegative reciprocal of the slope of the load line.

Also note that, because the area of the cross-hatched box (proportionalto RF power) decreases by a percentage equal to the percent decrease ofthe area of the solid box (proportional to DC power), a ratio of RFpower to DC power can remain constant. As mentioned previously,efficiency is generally proportional to the ratio of RF power to DCpower. As a result, even at differing power levels of the poweramplifier (as set based on number of active unit cells in the poweramplifier), efficiency of the power amplifier whose voltage and currentcharacteristics are shown in FIG. 8A can remain constant. As shown inFIG. 8B, another interpretation is that because voltage and currentcharacteristics for each individual unit cell that remains active canremain constant, efficiency of each individual unit cell that remainsactive also can remain constant. Therefore, efficiency of the poweramplifier whose voltage and current characteristics are shown in FIG. 8Acan remain constant even as number of unit cells that are active (orinactive) is varied during operation of the power amplifier.

According to several embodiments of the present disclosure, a scalableperiphery tunable matching (SPTM) power amplifier can achieve the twineffects shown in FIGS. 8A and 8B of selectively activating ordeactivating unit cells to adjust current (and therefore output power)while adjusting the load line corresponding to impedance seen by theentire power amplifier (i.e. N unit cells taken as a unit rather thanindividually) in such a manner that unit cells remaining active do notsee an individual change in output impedance and corresponding load lineslope. As used in the present disclosure, the scalable periphery (SP)aspect of the power amplifier refers to ability to activate ordeactivate unit cells within the power amplifier. The tunable matchingaspect of the power amplifier refers to ability to adjust the load linecorresponding to the output impedance seen by the power amplifier.Tunable matching networks are shown, for example, in U.S. Pat. No.7,795,968, which is incorporated herein by reference in its entirety.

FIG. 9A shows an exemplary SPTM power amplifier architecture (950). Asimilar architecture can also be found in U.S. Pat. No. 7,170,341,issued on Jan. 30, 2007 and entitled “Low Power Consumption AdaptivePower Amplifier”, which is incorporated herein by reference in itsentirety. By way of further example, and not of limitation, the SPTMpower amplifier of FIG. 9A can further comprise a tunable matchingnetwork (920) connected to the output of the scalable peripheryamplifier (910). Such a tunable matching network (920) can be configuredto dynamically adjust the output impedance seen by the SP amplifier(910), increasing the output impedance (adjusting load line forshallower slope) when unit cells are deactivated or decreasing theoutput impedance (adjusting load line for steeper slope) when more unitcells are activated. The SPTM power amplifier architecture (950) of FIG.9A is adapted to be connectable between a first terminal (930) and asecond terminal (935). The SPTM power amplifier architecture (950)comprises a driver (905) configured to receive and process a signal fromthe first terminal (930); a scalable periphery (SP) amplifier (910)connected with the driver (905) and configured to amplify a signaloutput from the driver (905); and a tunable matching network (920)connected with the SP amplifier (910) and configured to dynamicallyadjust output impedance seen by the SP amplifier (910). The SP amplifier(910) and the tunable matching network (920) together form what isreferred to as an SPTM power amplifier (900). Although the SPTM poweramplifier (900) may be adapted to amplify an output signal from a driver(905), as shown in FIG. 9A, the driver (905) is optional and may beremoved from FIG. 9A. As used in the present disclosure, a driver ordriver circuit may refer to an amplifier that precedes and drives (e.g.provides an output signal to) another amplifier. The person skilled inthe art will understand that for added flexibility, the driver may alsobe configured as an SPTM amplifier similar in architecture to (900).

According to several embodiments of the present disclosure, the SPamplifier (910) can comprise one or more unit cells operating inparallel. Each of the unit cells can comprise one or more active devices(e.g. transistors) that can be configured to operate as any class ofamplifier (e.g. class A, class B, class AB, class C, class D, class E,class F, etc.). The number of unit cells to be activated or deactivatedcan be determined as follows. Given an input power (e.g. usuallymeasured in milliwatts) provided to an SP amplifier and a desired powergain (e.g. usually measured in dB), an expected output power of the SPamplifier can be calculated according to techniques known to a personskilled in the art. Because the output power is proportional to an areaof the cross-hatched regions as illustrated in previous figures (e.g.FIG. 3), the expected output power can be used to determine a necessaryDC bias current. Based on the necessary DC bias current, a minimumnumber of unit cells capable of delivering the necessary DC bias currentcan be determined and in turn the number of unit cells to be turned ONcan be determined based on this minimum.

For the exemplary architecture (950) shown in FIG. 9A, the SP amplifier(910) can be constructed with any number of unit cells operating inparallel. The SP amplifier (910) can be constructed in a manner so as toallow each unit cell to be switched ON or OFF individually. In manyembodiments of the present disclosure, the switching circuitry can beconstructed using CMOS technology and various architectures known to theskilled person, such as, for example, the architecture presented in U.S.Pat. No. 7,910,993, issued on Mar. 22, 2011 and entitled “Method andApparatus for use in Improving Linearity of MOSFET's using anAccumulated Charge Sink”, and in U.S. Pat. No. 6,804,502, issued on Oct.12, 2004 and entitled “Switch Circuit and Method of Switching RadioFrequency Signals”, both of which are incorporated herein by referencein their entirety. Individual unit cells can be constructed using CMOS,silicon germanium (SiGe), gallium arsenide (GaAs), gallium nitride(GaN), bipolar transistors, or any other viable semiconductor technologyand architecture known, including one described in U.S. Pat. No.7,910,993, to or which can be envisioned by a person skilled in the art.Total DC bias current and current output of the SP amplifier (910) canbe adjusted by selectively switching ON or OFF individual unit cells. Insome embodiments of the present disclosure, unit cells can be arrangedaround a periphery of a selected area of a circuit.

FIG. 9B shows an SP amplifier with the tunable matching circuit at theinput. While this approach may not provide all of the benefits ofchanging the load line of the amplifier, it does help keep the inputimpedance constant as the number of active cells changes. Note that SPwithout TM can provide benefits above those of a standard amplifier.

FIG. 9C shows a combination of an SP amplifier with TM at the input andoutput. Such embodiment offers all the SPTM benefits plus adding abenefit of improved input matching. The tunable match not only adjustsload lines and compensates for changes in the SP amplifier, it is alsoused to adjust the frequency response.

If it is assumed that the unit cells of the SP amplifier (910) arenumbered 1 through N, where N is the total number of unit cells,activation and deactivation can proceed in a deterministic manner. If afirst power level requires that 5 unit cells be active, unit cells 1through 5 can be turned ON while other unit cells remain OFF. If asecond power level requires that 10 unit cells be active, unit cells 1through 10 can be turned ON while other unit cells remain OFF. FIG. 10Ashows an exemplary arrangement of unit cells (1010, 1020, 1030) of thescalable periphery amplifier (910) shown in FIG. 9A. A control wordcomprising one or more enable signals can be utilized to selectivelyactivate or deactivate the unit cells (1010, 1020, 1030). In a casewhere each unit cell (1010, 1020, 1030) can be controlled independentlyof one another, each unit cell (1010, 1020, 1030) can be selectivelyactivated or deactivated by a corresponding enable signal (1005) appliedto the unit cell (1010, 1020, 1030). Each unit cell (1010, 1020, 1030)can comprise one or more active devices that can be configured tooperate as any class of amplifiers. In the particular examplearrangement shown in FIG. 10A, each unit cell (1010, 1020, 1030)comprises a stack of two or more transistors (1015, 1017), although insome embodiments a unit cell may comprise a single transistor. A supplyvoltage (1050) can be connected to each unit cell (1010, 1020, 1030)through an inductor (1025), where the inductor (1025) is commonlyreferred to as an RF choke. A source terminal of the first transistor(1015) can be connected to ground (1060).

According to several embodiments of the present disclosure, the unitcells (1010, 1020, 1030) can be designed and constructed to haveidentical characteristics (e.g. transconductance, bias voltages,physical size, and so on). Furthermore, the unit cells (1010, 1020,1030) can be biased identically. With reference to FIG. 10A, a first FET(1015) of each unit cell (1010, 1020, 1030) can be biased identicallyand a second FET (1017) of each unit cell (1010, 1020, 1030) can bebiased identically (and differently from the first FET).

In alternative embodiments, the unit cells (1010, 1020, 1030) may bebiased differently and/or may be constructed to have differentcharacteristics (e.g. transconductance, bias voltages, physical size,and so on). For example, consider an SP amplifier that comprises a firstset of two unit cells, each constructed identically and biased at 20 mAof DC current, and a second set of four other unit cells, eachconstructed identically and biased at 10 mA of DC current. An outputpower level requiring 40 mA of DC bias current can be enabled byselectively activating either each unit cell in the first set of twounit cells or each unit cell in the second set of four unit cells.Alternatively, with reference to the same example, the same output powerlevel requiring 40 mA of DC bias current can be enabled by selectivelyactivating one unit cell from the first set of unit cells in conjunctionwith two unit cells from the second set of unit cells.

With reference back to FIG. 10A, the circuit shown in FIG. 10A comprisesa first terminal (1040) coupled to an optional driver (905) and a secondterminal (1045) coupled to the SP amplifier (910), where the firstterminal (1040) is adapted to receive an input signal to be amplified bythe SP amplifier (910) and the second terminal (1045) is adapted toreceive an output signal of the SP amplifier (910). An output of thedriver (905) at a node (1001) can be fed to an input node (1004) of theSP amplifier (910). An input signal at the input node (1004) of the SPamplifier (910) is in turn fed to each unit cell (1010, 1020, 1030) ofthe SP amplifier through an input coupling capacitor (1002) at the inputof each unit cell. The input coupling capacitor (1002) allows an RFsignal to pass from node (1001) to the input of each unit cell of the SPamplifier (910) while blocking a DC component of the input signal atnode (1040). The input coupling capacitor (1002) of each unit cell ofthe SP amplifier (910) is coupled to a gate of a first transistor (1015)in the stack of each unit cell (1010, 1020, 1030), while an outputsignal may be taken from a last (e.g. transistor farthest from the firsttransistor) transistor in the stack of each unit cell (1010, 1020,1030). In the particular case where the unit cell comprises a singletransistor, then a corresponding output signal is taken from the sametransistor to which the input signal is applied.

A bias network (1007) is connected to the bias input node (1006) of theSP amplifier (910), which is in turn routed to a node (1008) of eachunit cell connected to the gate of the first (e.g. input) transistor ofthe stack. The bias network (1007) can be a standard bias network thatwould be known to a person skilled in the art and, although notexplicitly shown in FIG. 10A, the bias network (1007) can supply biasvoltages to gates of transistors above the first transistor (1015) inthe stack, such as supply a bias voltage to the gate (1012) oftransistor (1017) in FIG. 10A. Alternatively, node (1012) can beconnected to a separate bias network (not shown) and supply a biasvoltage to transistor (1017) via this separate bias network. Gate biasvoltages of FETs generally have a direct effect on bias current. If theunit cell (e.g. 1010) comprises more than two transistors, gates ofthese other transistors can also be connected to bias networks asappropriate. The source of transistor (1015) can be connected to ground(1060) as shown in FIG. 10A or other circuits, which will be understoodby those skilled in the art.

It is to be noted that the skilled person will know other stackedamplification structures and may use these in lieu of the one suggestedin FIG. 10A and adapt the method herewith described to create an SPamplifier to such structures. FIG. 10F represents such a structure,wherein the stack is composed of P-type and N-type MOSFET devices in apush-pull configuration. In this configuration the top of the stack maycomprise a number of P-type MOSFET devices connected in a seriesconfiguration, and the bottom may comprise the same number of N-typeMOSFET devices also connected in a series configuration. The middle twodevices are thus of opposite types and interconnected at their drainterminals. In this embodiment the input RF is fed to the top and bottomdevices which in turn propagate the input through the top and bottomhalves of the stack in a complementary fashion, yielding in outputtingthe amplified RF output signal from one half of the stack or the otherhalf of the stack. In some embodiments both halves may outputsimultaneously but at different power levels. In some other embodimentsoutput symmetry of the push-pull arrangement can be modified byswitching off a number of P-type or N-type devices of the stack. Biasingof the gates may be provided via biasing terminals (1012, 1013). Theperson skilled in the art will understand that each of the transistors(1015, 1016, 1017, 1018) of FIG. 10F may be replaced by a cascodeconfiguration for higher RF output power capability. Cascode amplifiersrefer to common gate amplifier stages that are placed at the output of acommon source amplifier. Cascode stages increase gain by increasing theoutput resistance, improve the frequency response by mitigating theMiller capacitance effect in the common source stage, and can improvevoltage handling by dividing the voltage across the common gate andcommon source stages. Similarly, the person skilled in the art mayenvision a differential implementation of this embodiment, usingtransistors, cascode stages or even push-pull arrangements as activeelements of the unit cells.

While placing the input coupling capacitor (1002) within each unit cellmay have the advantage of using smaller capacitor sizes (one for eachunit), in some embodiments it may be desirable to place a single commoninput coupling capacitor outside of the unit cells as depicted by FIG.10G, wherein a single input coupling capacitor is placed outside the SPamplifier (910). In such a configuration, bias input node (1006) and SPinput node (1004) can be common, as depicted by FIG. 10G, oralternatively and for added flexibility in providing differing biases tothe input transistors (1015) of each unit cell, these nodes may beseparated as shown in FIG. 10A (thus routing the node (1006) of FIG. 10Gto node (1008) in lieu of node (1004)).

FIGS. 10B and 10C show embodiments of a unit cell of the scalableperiphery amplifier (910) of FIG. 9A. In these embodiments, anindividual unit cell can comprise a stack of transistors (1015, 1017).In the embodiment shown in FIG. 10B, an individual unit cell comprises aswitch (1070) whose state is controlled by a control signal (1005). Whenan individual unit cell is ON, the switch (1070) can couple an RF signaland a bias signal to the gate of the first transistor (1015). When anindividual unit cell is OFF, the switch (1070) can couple the gate ofthe first transistor (1015) to ground. Consequently, an individual unitcell can be turned OFF by cutting RF drive to and grounding the gate ofthe first transistor in the stack via the switch (1070), or vice versain the case that the individual unit cell is being turned ON. As alreadymentioned above, the stack may comprise a single transistor, in whichcase the first transistor is the only transistor of the stack.

In the embodiment shown in FIG. 10C, an individual unit cell can beturned OFF by grounding the gate of a second transistor (1017) in thestack via a switch (1080), which is not in the path of the RF signal. Toturn ON the individual unit cell, the switch (1080) can be operated toconnect the gate of the second transistor (1017) in the stack to a DCbias voltage applied at node (1024). In other embodiments, the stack canbe turned OFF by grounding (e.g. via a switch) higher (e.g. third orabove) FETs in the stack, thus reducing capacitance. This can result inhigher stress on devices in the stack due to unequal voltage division.

It is desirable to control the stress on the individual devices in thestack due to unequal voltage division across the devices (e.g. V_(DS) ofeach device) which may subject any one of the devices to a voltage closeto or larger than its limit breakdown voltage. As such, in someembodiments, the gates of the devices in the stack, with the exceptionof the input device (e.g. bottom device of FIG. 10A), are configured tofloat via insertion of a gate capacitor (1027, 1028) as depicted in FIG.10E. The value of the gate capacitor is chosen so to allow the gatevoltage to vary along (float) with the RF signal at the drain of thecorresponding device, which consequently allows control of the voltagedrop across the corresponding device, thus controlling the conduction ofthe device in accordance to the voltage at its drain, for a moreefficient operation of the device. In the case where this floatingtechnique is applied to several stacked devices, voltage across thedevices can be equalized by choosing the correct combination of gatecapacitor and gate bias voltage for each of the devices. The skilledperson will appreciate the difference of such a gate capacitor, as it isnot used as a traditional bypass capacitor. Teachings about thisfloating technique, also referred to as conduction controlling circuit,can be found in U.S. Pat. No. 7,248,120, which is incorporated herein byreference in its entirety.

With reference to FIG. 10E, when the switch (1012) grounds the gate ofdevice (1017), this device cannot operate in the floating mode via gatecapacitor (1028) and as such will see a larger voltage (e.g. twice aslarge) applied across its drain and source terminals as the voltageequalization chain is broken. In order to protect this device (1017)from excessive voltage, in one embodiment the bias voltage applied tothe other devices in the stack (above (1017)) may be modified tore-equalize the voltage across these devices which consequently lowersthe voltage across (1017). A switch (e.g. (1080 a)) may be associated toeach device used in the voltage equalization chain to select from one oftwo DC bias voltages (DC bias1, DC bias2), one for the case when device(1017) is part of the equalization chain and one for the case when thedevice (1017) is not part of the equalization chain. The same enablesignal (1005) used to activate/deactivate the unit cell may be used tocontrol the switch (1080 a).

Switching the input device will change the impedance looking into theamplifier core. Use of a tunable input matching circuit can compensatefor these changes. The use of techniques that switch one of the cascodedevices (a device higher in the stack) will have a much smaller effecton the amplifier's input impedance. This is because the gate bias on thebottom or common source stage will be maintained and thus the change incapacitance will be reduced.

FIG. 10D shows an embodiment of a unit cell of the SP amplifier (910) ofFIGS. 9A-9B that allows variable bias current in the unit cell. A personskilled in the art may recognize the arrangement shown in FIG. 10D as acurrent mirror. A reference current (1043) flowing through transistors(1035, 1037) will be approximately copied as a bias current in thetransistors (1015, 1017) that serve to amplify an RF input signalapplied to the input node (1008). A bias voltage, not shown, is appliedto the gates of the cascode transistors (1037) (1017). Consequently,changing the reference current (1043) will result in changing the biascurrent flowing through the transistors (1015, 1017) and consequentlycontrol the ON or OFF state of the unit cell as well. Switches similarto switches (1070, 1080) employed in FIGS. 10B and 10C can also beutilized in the unit cell of FIG. 10D to control ON or OFF state of theunit cell. A person skilled in the art will appreciate that variouscombinations of the embodiments shown in FIGS. 10A-10D are also possibleand are within the scope of the present disclosure.

The switches (1070, 1080) shown in FIGS. 10B and 10C used for turning ONor OFF the unit cell can be operated by enable signals (1005). Suchenable signals (1005) may be generated by control circuitry within an SPor SPTM amplifier itself, a transmitter that drives the SP or SPTMamplifier, outside control circuitry (e.g. a controller of a cellularphone), or some other source. The switches (1070, 1080) can beconstructed as transmission gates, by using FETs, by using BJTs, or byany other means for implementing a switch that are known to or that canbe envisioned by a person skilled in the art (e.g. U.S. Pat. No.7,910,993 previously mentioned).

Furthermore, as previously mentioned, given an input power (e.g. usuallymeasured in milliwatts) provided to an SP or SPTM amplifier and adesired power gain (e.g. usually measured in dB), an expected outputpower of the SP or SPTM amplifier can be calculated according totechniques known to a person skilled in the art. Because the outputpower is proportional to an area of the cross-hatched regions asillustrated in previous figures (e.g. FIG. 3), the expected output powercan be used to determine a necessary DC bias current. Based on thenecessary DC bias current, a minimum number of unit cells capable ofdelivering the necessary DC bias current can be determined and in turnthe number of unit cells to be turned ON can be determined based on thisminimum. A control word comprising one or more control signals (1005)can be generated to control ON or OFF states of the unit cells based onthe above calculations/determinations.

According to the method of reducing bias current in combination withincreasing load impedance seen by the entire power amplifier (previouslydescribed with reference to current and voltage graphs shown in FIGS. 8Aand 8B), inactive unit cells can be activated to increase total outputcurrent of the SP amplifier (910) or active unit cells can bedeactivated to decrease total output current of the SP amplifier (910),as needed, to adjust for differing input signal power levels. Controlcircuitry comprised of a controller (not shown) and signal linesconnecting the controller to individual unit cells can selectivelyactivate or deactivate the individual unit cells. By way of example andnot of limitation, the controller may be a portion of a control unit ofa cellular phone or wireless device. A person skilled in the art willrealize that an amplifier can be constructed using a scalable peripheryarchitecture alone, without tunable matching. Also scalable peripheryarchitecture alone can achieve significant benefits, as alreadymentioned above.

With reference back to FIGS. 7A and 7B, a scalable peripheryarchitecture alone can be controlled to reduce DC bias current andthereby reduce DC power dissipation. As previously mentioned, reduced DCpower dissipation with reduced RF signal input drive can result inhigher drain efficiency than a fixed amplifier whose DC powerdissipation is constant (see FIGS. 3 and 4). With reference to FIGS. 3and 4, for a fixed amplifier, when the RF drive decreases, the RF outputpower decreases even as DC power dissipation remains fixed. Fixed DCpower dissipation in the graphs of FIGS. 3 and 4 is indicated by a fixedsize of the solid box. Reduced RF output power is indicated by a reducedsize of the cross-hatched box in FIG. 4 as compared to FIG. 3. Acombination of reduced RF output power with fixed DC power dissipationcan result in lower drain efficiency as RF drive decreases. In contrast,in the graphs of FIG. 7A, although RF output power is reduced withreduced RF drive, DC power dissipation is also decreased. Reduced drainefficiency for the case of reduced RF drive can be a salient issue inapplications such as voice communications, where the amplifier may beoperating at maximum power for a small percentage of total time.

With reference back to FIG. 9A, the SPTM power amplifier architecture(950) also comprises a tunable matching network (920) connected to theoutput of the SP amplifier (910). Such a tunable matching network (920)can be configured to dynamically adjust the output impedance seen by theSP amplifier (910), increasing the output impedance (adjusting the loadline for a shallower slope) when unit cells are deactivated ordecreasing the output impedance (adjusting the load line for a steeperslope) when more unit cells are activated. Through load line adjustmentby way of the tunable matching network (920), the individual unit cellsthat remain active can see a constant output impedance.

A person skilled in the art will be aware that a fixed (e.g.non-tunable) impedance matching network can be constructed usingdiscrete capacitors and inductors when a wavelength corresponding to afrequency of operation is large compared to the size of such capacitorsand inductors and parasitic effects of such capacitors and inductors arenot significant at the frequency of operation, or by using distributedelements (e.g. open circuit or short circuit stubs placed in series orin parallel between the load and output terminals of the poweramplifier) when the wavelength corresponding to the frequency ofoperation is small compared to the size of the circuit elements. Designprocedures and equations for fixed impedance matching networks can bereadily found in textbooks addressing the subjects of RF engineering andmicrowave engineering and are assumed to be known to a person skilled inthe art.

FIG. 11 shows an exemplary tunable matching network that can be utilizedin the SPTM power amplifier architecture (950) of FIG. 9A. The tunablematching network can comprise a tunable reactive element (1110) placedin series between two impedances to be matched and a tunable reactiveelement (1120) placed in parallel relative to two impedances to bematched. Details regarding tunable reactive elements, including tunablecapacitors and inductors, are described, for example, in PCT publicationnumber WO2009/108391 entitled “Method and Apparatus for use in DigitallyTuning a Capacitor in an Integrated Circuit Device”, published on Sep.3, 2009, and in U.S. patent application Ser. No. 13/595,893 entitled“Method and Apparatus for Use in Tuning Reactance in an IntegratedCircuit Device”, filed on Aug. 27, 2012, both incorporated by referenceherein in their entirety, where examples of digital tuning capacitorsand/or digital tuning inductors for use in a tunable matching networkare disclosed. A person skilled in the art will be aware that matchingnetworks such as shown in FIG. 11 can be joined together in a 7r-configuration, a t-configuration, or a cascade configuration as neededto achieve necessary bandwidth characteristics and quality factor (Qfactor) and can be configured to transform impedances upward ordownward. FIG. 19 shows an exemplary realization of the tunable matchingnetwork of FIG. 11, wherein a combination of switches and variablecapacitors and inductors are used.

A person skilled in the art will also be aware that such an SPTMarchitecture (e.g. as shown in FIGS. 9A-9C) may be constructed on asingle chip where both the SP and TM components are integratedmonolithically, or alternatively the TM component may be partiallyintegrated (or not at all). Various configurations and correspondingpartitioning within a single or multiple chips may depend on the usedcomponents and related technologies.

According to several embodiments of the present disclosure, withreference to FIG. 9A, operation of the SP amplifier (910) and the TMnetwork (920) is generally dependent on each other. In general,impedance presented by the TM network (920) depends on the number ofunit cells turned ON or OFF in the SP amplifier (910). Consequently,control signals generated by control circuitry associated with the SPamplifier (910) and those generated by control circuitry associated withthe TM network (920) can be dependent on one another. Control circuitryof the TM network (920) can be configured to adjust individual tunablereactive elements as needed to result in a desired value of impedancepresented by the TM network (920) to the output of the SP amplifier(910). In some embodiments, the control circuitry for both the SPamplifier (910) and the TM network (920), as well as the SP amplifieritself can be integrated in one control circuit using for examplesilicon technologies, which include, but are not limited to,complementary metal oxide semiconductor (CMOS) technology, silicon oninsulator (SOI) CMOS technology, silicon on sapphire (SOS) CMOStechnology, and bipolar CMOS (BiCMOS) technology (e.g. technologyinvolving a combination of bipolar junction transistors and CMOStransistors).

In addition to adjusting for differing signal input power levels, theSPTM architecture allows tuning of the amplifier for differingfrequencies of operation and/or differing wireless standards (e.g. LTE,WiMAX, or 4G, among others) and modes (modulation and thus bandwidth,PAR, . . . ) over the range of output powers, and thus reduce thetrade-off between linearity and efficiency previously discussed. Suchadjustment would comprise retuning (e.g. via control) the amplifier toachieve the desired impedance match and output power level at a givenfrequency of operation and/or for a given wireless standard/mode. Thisflexibility of tuning for specific parameters allows customization ofthe SPTM amplifier to customer requirements.

In one embodiment, SPTM can be put in a feed forward control system thatmonitors the power levels, PAR, frequency and other related operationalparameters of the SPTM. In another embodiment SPTM can be used in anopen loop manner. In open loop, SPTM settings as a function of power andfrequency would be calibrated and stored. A lookup table could thenconvert frequencies and powers into SPTM settings. Same principle ofusing lookup tables may also be applied to a closed loop configuration,whereby the system monitors the related operational parameters and usesthe content of the lookup tables to automatically provide correctionsand/or changes to the SPTM operation. The calibration table andlookup/conversion function can reside in the PA itself, or in thebaseband/transceiver IC (hardware and/or software). It is envisionedthat one could store calibration information in the PA using fuses,EEPROM, laser trim, or directly writing to the registers from a basebandIC. If the calibration information is stored in the PA itself, it can bepre-configured in IC production so that all PAs reaching the handsetmanufacturer look the same.

A person skilled in the art will realize that the scalable peripheryamplifier or scalable periphery tunable matching amplifier of thedisclosure can be used as an intermediate stage as well as a driverstage in an RF device. Furthermore, efficiency can be optimized bycombining multiple amplifiers in a signal chain and selectively routingthe signal through certain amplifiers depending on desired output power,such as described, for example, in U.S. Pat. No. 7,795,968, issued onSep. 14, 2010 and entitled “Power Ranging Transmit RF Power Amplifier,which is incorporated herein by reference in its entirety. In such aconfiguration, any or all of the multiple amplifiers may be designedusing the scalable periphery amplifier or scalable periphery tunablematching amplifier of the present disclosure.

FIG. 12 shows an exemplary embodiment of the disclosure comprisingmultiple SP(TM) amplifiers. The two amplifiers (1220) and (1250) canboth be SP (scalable periphery) amplifiers or SPTM (scalable peripherytunable matching) amplifiers or some combination thereof. Although theconfiguration shown in FIG. 12 only shows two amplifiers, a personskilled in the art will realize that more amplifiers can also beconnected in cascade or in parallel. An input signal applied to inputterminal (1210) can be amplified by SP(TM) amplifier (1220). A switch(1230) can then route the signal to a through circuit (1240) that isconfigured to pass the signal unaltered directly to an output terminal(1260) for lower output power or to a second SP(TM) amplifier (1250) forhigher output power. Consider an exemplary case where the amplifier(1220) comprises both a scalable periphery amplifier (910) and a tunablematching network (920) (e.g. as shown in FIG. 9A). Because the switch(1230) can be operated to direct the signal directly to the output(1260) or to the second SPTM amplifier (1250), the tunable matchingnetwork (920) of the SPTM amplifier (1220) can be useful becauseimpedance at the output (1260) may be different than an input impedanceof the second SP(TM) amplifier (1250), requiring impedance presented toan output of the SP(TM) amplifier (1220) to be different depending onconnection established by the switch (1230). In some embodiments theswitch (1230) may be omitted to provide direct coupling between the twoamplifiers (1220) and (1250). In some embodiments the amplifier (1220)may be a driver stage, and the amplifier (1250) a final stage.

As discussed previously, DC bias current to individual unit cells can bevaried to achieve certain desired ACLR vs. output power characteristics.FIG. 17 shows ACLR as a function of output power for both class A andclass AB operation, corresponding to the SPTM amplifier with a total ofN number of unit cells. With reference to the curve corresponding toclass AB operation with all unit cells operating, output power can beadjusted by adjusting input power. FIG. 16 shows PAE as a function ofoutput power for both class A and class AB, corresponding to the samedevice shown in FIG. 17, and FIG. 15 is a combined representation ofFIGS. 16 and 17.

A given design specification may require operation at or above a certainpower level (e.g. P_(spec)) and below a certain ACLR (e.g. ACLR_(spec))as shown in FIG. 14. Performance variation can result from, for example,changes in frequency, battery voltage, power level, and othercharacteristics. Normally, in order to ensure proper operation despitesuch variations, an amplifier is designed to provide margins for bothACLR (e.g. Margin_(ACLR)) and output power (e.g. Margin_(Power)). Asoutput power (e.g. P_(out)) changes in a conventional amplifier or anSPTM amplifier with all unit cells constantly active, an operating pointmoves along the curve. In some embodiments, the entire curve can beshifted to the left (e.g. by turning OFF unit cells) or to the right(e.g. by turning ON unit cells) rather than shifting an operating pointalong the curve. Shifting the entire curve left or right rather thanshifting an operating point along the curve allows a designer to reducethe margins (e.g. Margin_(ACLR) and Margin_(Power)) necessary to ensurethat amplifier operation remains within desired specifications. AlthoughACLR is used here as a reference to a desired operation of the amplifierfor a given output power, the skilled person will realize that otherreferences, such as linearity, drain current, out of band emissions, Rxband noise, EVM and the like may also be used individually or incombination with one another.

For example, in reference to FIG. 14, a point labeled P_(O1SPTM)corresponds to an output power level with half of the total number ofunit cells operating, and a point labeled P_(O2SPTM) corresponds to anoutput power level with all of the unit cells operating. Both pointsP_(O1SPTM) and P_(O2SPTM) correspond to operation in a class AB null(represented by a dip in the ACLR versus Pout curve). In someembodiments of the present disclosure, adjustment of final size shiftingthe entire curve left or right and/or switching the PA between differentclasses of operation (e.g. switching between class A and class AB)enables reduction of the margins needed to guarantee desired performanceover a range of frequencies, battery voltages, power levels, and othercharacteristics. Such adjustment may be obtained in a closed loop oropen loop control configuration, which may include lookup tables and/orcalibration tables containing mapping between amplifier's operatingcharacteristic and the various affecting parameters.

In a further embodiment, the SPTM amplifier shown in FIG. 9A can beadjusted during circuit operation to operate either as a class Aamplifier or a class AB amplifier. The scalable periphery amplifier(910) and/or the tunable matching network (920) can be adjusted tochange the SPTM amplifier from class A operation to class AB operationor vice versa. In a further embodiment, individual unit cells of thescalable periphery amplifier (910) can be individually adjusted tooperate either as a class A amplifier or a class AB amplifier. Forexample, one half of active unit cells may be adjusted to operate asclass A amplifiers while a remaining half of active unit cells may beadjusted to operate as class AB amplifiers.

The curves of FIG. 17 correspond to an exemplary case of 10 MHz LTEbaseband signal, 782 MHz carrier frequency, QPSK modulation and coding,115 mA of DC bias current, and a supply voltage of 3.5 V. In the curvesof FIG. 17, output power was adjusted by appropriate adjustment of inputpower.

Although class A operation can result in lower ACLR at output powersbelow approximately 26 dBm, at higher output powers class AB operationresults in lower ACLR. Thus, at output power levels below approximately26 dBm, class A operation can be selected, while at output power levelsabove approximately 26 dBm, class AB operation can be selected. Class Aand class AB operation exhibit different ACLR versus output powercharacteristics because of different linearities and differentconduction angles (a class A amplifier conducts current throughout anentire cycle of oscillation of the amplifier, while a class AB amplifierconducts current through between 50% and 100% of the entire cycle ofoscillation of the amplifier). The conduction angle generally refers toan angle measure corresponding to a percentage of a cycle of oscillationduring which the amplifier is conducting. A class A amplifier, forexample, may be stated to have a conduction angle of 2π radians or 360degrees because it conducts throughout the entire cycle of oscillation.

Appropriate adjustment of the SPTM amplifier can maintain operation in aclass AB null. Switching between class A operation and class ABoperation can be useful in keeping ACLR below a desired limit (e.g. −35dBc).

Different systems and different standards may have different ACLRrequirements that must be met. For example, many LTE systems may allow amaximum ACLR of −33 dBc. In general, ACLR requirements are determined bya physical distance between the amplifier and a receiver operating on anadjacent channel that may receive interference from the amplifier. Forexample, geographic layout of a cellular network as well as frequencyallocation can play a role in determining such distance. Manycommercially available amplifiers exhibit maximum ACLR of −35 to −37dBc.

The SPTM amplifier (910) (see, e.g., FIG. 10A) can be switched betweenclass A and class AB operation by adjusting a bias of one or more of thetransistors in a unit cell.

Furthermore, optimum PAE (power added efficiency) for a class AB linearPA occurs in the class AB null (represented by a dip in the ACLR versusPout curve, e.g. FIG. 17). By optimally choosing the SPTM switch points(primarily the SP switch points, e.g. adjusting the number of unit cellsthat are ON) as a function of output power, the PA can be kept in theclass AB null. As used herein, the term “SPTM switch points” may referto points at which final size of the scalable periphery amplifier (910)and/or the tunable matching network (920) (see, e.g., FIG. 9A) ischanged. For example, a first SPTM switch point may correspond to afirst power level (e.g. 15 dBm), where dropping below the first powerlevel may correspond to changing a number of active unit cells from afirst number (e.g. 11 unit cells) to a second number (e.g. 10 unitcells) and increasing above the first power level may correspond tochanging the number of active unit cells from the second number to thefirst number.

Choosing of optimal SPTM switch points generally involves accurateknowledge of the target output power, as trying to get too much powerout of a certain final size can result in ACLR that is above a givenspecification. For an exemplary case of cell phone communications, somefactors that are used to determine target output power include distancefrom a cell phone to a nearest base station (e.g. greater distance mayindicate higher target output power) as well as whether the cell phoneis transmitting data or voice (e.g. data transmission generally requireshigher power than voice transmission). Based on such factors, thenearest base station may signal the cell phone indicating the targetoutput power that the cell phone should use when transmitting.

As discussed previously, turning OFF unit cells can shift the entirecurve of FIG. 17 to the left, while turning ON unit cells can shift theentire curve to the right. As a result, operation of the SPTM amplifiercan be maintained in the AB null at different power levels. Control ofthe SPTM amplifier can be implemented as either open loop (e.g. nofeedback) or closed loop (including feedback) control. It may also behelpful to adjust bias voltage in addition to selectively activating ordeactivating unit cells. FIGS. 13 and 14 show shifting of the ACLRversus output power curves when half of the unit cells are turned offfor class A and class AB operations respectively.

Alternatively, use of switching to include or omit harmonic terminations(e.g. the harmonic shorts or opens) in conjunction with high RF inputcan adjust an individual unit cell to operate as a switching amplifier(e.g. class D, E, F). For example, in order to operate as a class Eamplifier, a capacitor can be connected to the drain. In order tooperate as a class F amplifier, even harmonics can be shorted and oddharmonics can be opened. Furthermore, in some embodiments harmonictermination can be used to further “tweak” a response of the amplifierto obtain a desired operating characteristic (e.g. linearity, ACLR,output power, drain current, out of band emissions, Rx band noise, EVM,and so forth) within a given class of operation.

By way of example and not of limitation, FIG. 18 shows an exemplarymethod of switching between the different classes (e.g., class A, AB, B,C, E, F) of amplifier operation. The scalable periphery amplifier isconnected with an output tunable matching circuit, and the tunablematching circuit can comprise a switch that selects the desired class ofoperation (e.g. harmonic termination). The variable impedance matchingnetwork can change the impedance to match the impedance of the scalableperiphery amplifiers with the load, depending on the class of amplifieroperation that is selected. In some embodiments usage of switches toselect a harmonic termination for a desired class of operation and asdepicted by FIG. 18 may be avoided. FIG. 20 shows such an embodiment,wherein instead of selecting fixed impedances using a switch, variableimpedance circuits (2010, 2020) are used to provide the desired harmonictermination (f₀ denoting a fundamental frequency, typically the centeroperating frequency of the amplifier, and n.f₀ the correspondingharmonics) to the output of the SP amplifier. This configuration of FIG.20 has the added advantage to remove any energy dissipation due to theswitches of FIG. 18 which are used to switch a path to the RF outputsignal from the amplifier. It is to be noted that switches may be usedwithin the variable impedance circuits (2010, 2020) of FIG. 20.

The examples set forth above are provided to give those of ordinaryskill in the art a complete disclosure and description of how to makeand use the embodiments of the scalable periphery tunable matching poweramplifier of the disclosure, and are not intended to limit the scope ofwhat the inventors regard as their disclosure.

Modifications of the above-described modes for carrying out the methodsand systems herein disclosed that are obvious to persons of skill in theart are intended to be within the scope of the following claims. Allpatents and publications mentioned in the specification are indicativeof the levels of skill of those skilled in the art to which thedisclosure pertains. All references cited in this disclosure areincorporated by reference to the same extent as if each reference hadbeen incorporated by reference in its entirety individually.

It is to be understood that the disclosure is not limited to particularmethods or systems, which can, of course, vary. It is also to beunderstood that the terminology used herein is for the purpose ofdescribing particular embodiments only, and is not intended to belimiting. As used in this specification and the appended claims, thesingular forms “a”, “an”, and “the” include plural referents unless thecontent clearly dictates otherwise. The term “plurality” includes two ormore referents unless the content clearly dictates otherwise. Unlessdefined otherwise, all technical and scientific terms used herein havethe same meaning as commonly understood by one of ordinary skill in theart to which the disclosure pertains.

A number of embodiments of the disclosure have been described.Nevertheless, it will be understood that various modifications can bemade without departing from the spirit and scope of the presentdisclosure. Accordingly, other embodiments are within the scope of thefollowing claims.

The invention claimed is:
 1. An amplification circuit, comprising: oneor more amplifiers configured to be selectively activated ordeactivated, wherein each amplifier of the one or more amplifierscomprises: a stack of a plurality of transistors, and one or more gatecapacitors connected to respective one or more transistors of theplurality of transistors; wherein, in each amplifier: an inputtransistor of the plurality of transistors is configured to receive aninput signal; a cascode transistor of the plurality of transistors isconfigured to receive a cascode gate voltage held to a constant DC levelregardless of the amplifier being active or inactive; the one or moregate capacitors are connected between one or more gates of therespective one or more transistors and a reference ground with theexception of the input transistor, and a non-bypassing gate capacitor ofthe one or more gate capacitors is configured to allow a gate voltage ofa respective transistor of the plurality of transistors to vary alongwith a radio frequency (RF) voltage at a drain of the respectivetransistor.
 2. The amplification circuit of claim 1, wherein each of theone or more amplifiers comprises a switch connected in series with agate of the input transistor, the switch being configured to open orclose the input signal path to the input transistor, thus selectivelyactivating or deactivating a corresponding amplifier.
 3. Theamplification circuit of claim 2, wherein the switch is configured toselectively connect the gate of the input transistor to either the inputsignal or to a fixed voltage, so as to selectively activate ordeactivate the one or more amplifiers.
 4. The amplification circuit ofclaim 1, wherein each amplifier of the one or more amplifiers is abiased amplifier, a bias being applied to the biased amplifier.
 5. Theamplification circuit of claim 2, wherein each switch is controlled by acontrol circuitry of the amplification circuit.
 6. The amplificationcircuit of claim 1, further comprising an output tunable matchingnetwork operatively connected to an output of the amplification circuit,wherein the tunable matching network is configured to adjust an outputload impedance seen by the output of the amplification circuit.
 7. Theamplification circuit of claim 6, wherein the output tunable matchingnetwork comprises: one or more tunable reactive elements connectedbetween the output of the amplification circuit and the output load; anda tunable matching control circuit configured to tune the one or moretunable reactive elements to adjust the output load impedance seen bythe output of the amplification circuit.
 8. The amplification circuit ofclaim 6, wherein the output tunable matching network comprises: a firstset of one or more tunable reactive elements placed in series betweenthe output of the amplification circuit and the output load impedance;and/or a second set of one or more tunable reactive elements placed inparallel with the output load impedance, and a tunable matching controlcircuitry configured to tune the first set of the one or more tunablereactive elements and/or the second set of the one or more tunablereactive elements, thus adjusting the output load impedance seen by theoutput of the amplification circuit.
 9. The amplification circuit ofclaim 6, wherein the output tunable matching network comprises aplurality of tunable matching networks arranged in one or more of a: a)π-configuration, b) t-configuration, and c) cascade configuration. 10.The amplification circuit of claim 8, wherein the one or more tunablereactive elements comprise one or more of: a) one or more digital tuningcapacitors, and b) one or more digital tuning inductors.
 11. Theamplification circuit of claim 8, wherein the one or more amplifiers andone or more of: a) the output tunable matching network in entirety or inpart, and b) the tunable matching control circuitry in entirety or inpart, are monolithically integrated.
 12. The amplification circuit ofclaim 1, further comprising a harmonic termination network connected toan output of the amplification circuit.
 13. The amplification circuit ofclaim 12, herein the harmonic termination network comprises one or moretunable reactive elements.
 14. The amplification circuit of claim 13,wherein the one or more tunable reactive elements are connected in oneof a) series between the output of the amplification circuit and anoutput load or b) in parallel at the output of the amplificationcircuit, or a combination thereof.